Method and system for single weight antenna system for hsdpa

ABSTRACT

In a RF communications system, aspects for single weight antenna system for HSDPA may comprise receiving HSDPA signals via a plurality of receive antennas and individually adjusting a phase of a portion of the received HSDPA signals via a single weight. The phase adjusted portion of the received HSDPA signals may be combined with at least one of the received HSDPA signals to generate combined HSDPA signals. At least one control signal may control the adjusting of the phase of the received HSDPA signals. Discrete phases may be communicated to adjust the phase of the portion of the received HSDPA signals, where the plurality of the discrete phases may range from zero radians to substantially 2π radians. Phase shift channel estimates may be generated during the identified time to determine the discrete phase. A desired phase may be generated from the phase shift channel estimates, and the single weight may be generated from the desired phase.

CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE

This application makes reference, claims priority to, and claims thebenefit of U.S. Provisional Application Ser. No. 60/616,686 filed Oct.6, 2004.

The present application is related to the following applications, eachof which is incorporated herein by reference in its entirety for allpurposes:

-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16199US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16200US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16201US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16202US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16203US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16204US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16205US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16206US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16207US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16208US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16209US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16210US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16211US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16213US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16214US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16215US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16216US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16217US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16218US02) filed Jun. 30, 2005;-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16219US02) filed Jun. 30, 2005; and-   U.S. patent application Ser. No. ______ (Attorney Docket No.    16220US02) filed Jun. 30, 2005.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to receiving radio frequencysignals. More specifically, certain embodiments of the invention relateto a method and system for single weight antenna system for HSDPA.

BACKGROUND OF THE INVENTION

Mobile communication has changed the way people communicate and mobilephones have been transformed from a luxury item to an essential part ofevery day life. The use of mobile phones is today dictated by socialsituations, rather than hampered by location or technology. While voiceconnections fulfill the basic need to communicate, and mobile voiceconnections continue to filter even further into the fabric of every daylife, the mobile Internet is the next step in the mobile communicationrevolution. The mobile Internet is poised to become a common source ofeveryday information, and easy, versatile mobile access to this datawill be taken for granted.

Third generation (3G) cellular networks have been specifically designedto fulfill these future demands of the mobile Internet. As theseservices grow in popularity and usage, factors such as cost efficientoptimization of network capacity and quality of service (QoS) willbecome even more essential to cellular operators than it is today. Thesefactors may be achieved with careful network planning and operation,improvements in transmission methods, and advances in receivertechniques. To this end, carriers need technologies that will allow themto increase downlink throughput and, in turn, offer advanced QoScapabilities and speeds that rival those delivered by cable modem and/orDSL service providers. In this regard, networks based on wideband CDMA(WCDMA) technology may make the delivery of data to end users a morefeasible option for today's wireless carriers.

FIG. 1 a is a technology timeline indicating evolution of existing WCDMAspecification to provide increased downlink throughput. Referring toFIG. 1 a, there is shown data rate spaces occupied by various wirelesstechnologies, including General Packet Radio Service (GPRS) 100,Enhanced Data rates for GSM (Global System for Mobile communications)Evolution (EDGE) 102, Universal Mobile Telecommunications System (UMTS)104, and High Speed Downlink Packet Access (HSDPA) 106.

The GPRS and EDGE technologies may be utilized for enhancing the datathroughput of present second generation (2G) systems such as GSM. TheGSM technology may support data rates of up to 14.4 kilobits per second(Kbps), while the GPRS technology, introduced in 2001, may support datarates of up to 115 Kbps by allowing up to 8 data time slots per timedivision multiple access (TDMA) frame. The GSM technology, by contrast,may allow one data time slot per TDMA frame. The EDGE technology,introduced in 2003, may support data rates of up to 384 Kbps. The EDGEtechnology may utilizes 8 phase shift keying (8-PSK) modulation forproviding higher data rates than those that may be achieved by GPRStechnology. The GPRS and EDGE technologies may be referred to as “2.5G”technologies.

The UMTS technology, introduced in 2003, with theoretical data rates ashigh as 2 Mbps, is an adaptation of the WCDMA 3G system by GSM. Onereason for the high data rates that may be achieved by UMTS technologystems from the 5 MHz WCDMA channel bandwidths versus the 200 KHz GSMchannel bandwidths. The HSDPA technology is an Internet protocol (IP)based service, oriented for data communications, which adapts WCDMA tosupport data transfer rates on the order of 10 megabits per second(Mbits/s). Developed by the 3G Partnership Project (3GPP) group, theHSDPA technology achieves higher data rates through a plurality ofmethods. For example, many transmission decisions may be made at thebase station level, which is much closer to the user equipment asopposed to being made at a mobile switching center or office. These mayinclude decisions about the scheduling of data to be transmitted, whendata is to be retransmitted, and assessments about the quality of thetransmission channel. The HSDPA technology may also utilize variablecoding rates. The HSDPA technology may also support 16-level quadratureamplitude modulation (16-QAM) over a high-speed downlink shared channel(HS-DSCH), which permits a plurality of users to share an air interfacechannel

In some instances, HSDPA may provide a two-fold improvement in networkcapacity as well as data speeds up to five times (over 10 Mbit/s) higherthan those in even the most advanced 3G networks. HSDPA may also shortenthe roundtrip time between network and terminal, while reducingvariances in downlink transmission delay. These performance advances maytranslate directly into improved network performance and highersubscriber satisfaction. Since HSDPA is an extension of the GSM family,it also builds directly on the economies of scale offered by the world'smost popular mobile technology. HSDPA may offer breakthrough advances inWCDMA network packet data capacity, enhanced spectral and radio accessnetworks (RAN) hardware efficiencies, and streamlined networkimplementations. Those improvements may directly translate into lowercost-per-bit, faster and more available services, and a network that ispositioned to compete more effectively in the data-centric markets ofthe future.

The capacity, quality and cost/performance advantages of HSDPA yieldmeasurable benefits for network operators, and, in turn, theirsubscribers. For operators, this backwards-compatible upgrade to currentWCDMA networks is a logical and cost-efficient next step in networkevolution. When deployed, HSDPA may co-exist on the same carrier as thecurrent WCDMA Release 99 services, allowing operators to introducegreater capacity and higher data speeds into existing WCDMA networks.Operators may leverage this solution to support a considerably highernumber of high data rate users on a single radio carrier. HSDPA makestrue mass-market mobile IP multimedia possible and will drive theconsumption of data-heavy services while at the same time reducing thecost-per-bit of service delivery, thus boosting both revenue andbottom-line network profits. For data-hungry mobile subscribers, theperformance advantages of HSDPA may translate into shorter serviceresponse times, less delay and faster perceived connections. Users mayalso download packet-data over HSDPA while conducting a simultaneousspeech call.

HSDPA may provide a number of significant performance improvements whencompared to previous or alternative technologies. For example, HSDPAextends the WCDMA bit rates up to 10 Mbps, achieving higher theoreticalpeak rates with higher-order modulation (16-QAM) and with adaptivecoding and modulation schemes. The maximum QPSK bit rate is 5.3 Mbit/sand 10.7 Mbit/s with 16-QAM. Theoretical bit rates of up to 14.4 Mbit/smay be achieved with no channel coding. The terminal capability classesrange from 900 kbit/s to 1.8 Mbit/s with QPSK modulation, and 3.6 Mbit/sand up with 16-QAM modulation. The highest capability class supports themaximum theoretical bit rate of 14.4 Mbit/s.

However, implementing advanced wireless technologies such as WCDMAand/or HSDPA may still require overcoming some architectural hurdles.For example, the RAKE receiver is the most commonly used receiver inCDMA systems, mainly due to its simplicity and reasonable performanceand WCDMA Release 99 networks are designed so that RAKE receivers may beused. A RAKE receiver contains a bank of spreading sequence correlators,each receiving an individual multipath signal. A RAKE receiver operateson multiple discrete paths. The received multipath signals can becombined in several ways, from which maximal ratio combining (MRC) ispreferred in a coherent receiver. However, a RAKE receiver may besuboptimal in many practical systems, for example, its performance maydegrade from multiple access interference (MAI), that is, interferenceinduced by other users in the network.

The utilization of multiple transmit and/or receive antennas is designedto introduce a diversity gain and to suppress interference generatedwithin the signal reception process. Such diversity gains improve systemperformance by increasing received signal-to-noise ratio, by providingmore robustness against signal interference, and/or by permittinggreater frequency reuse for higher capacity. In communication systemsthat incorporate multi-antenna receivers, a set of M receive antennasmay be utilized to null the effect of (M−1) interferers, for example.Accordingly, N signals may be simultaneously transmitted in the samebandwidth using N transmit antennas, with the transmitted signal thenbeing separated into N respective signals by way of a set of N antennasdeployed at the receiver. Systems that utilize multiple transmit andreceive antennas may be referred to as multiple-input multiple-output(MIMO) systems. One attractive aspect of multi-antenna systems, inparticular MIMO systems, is the significant increase in system capacitythat may be achieved by utilizing these transmission configurations. Fora fixed overall transmitted power, the capacity offered by a MIMOconfiguration may scale with the increased signal-to-noise ratio (SNR).For example, in the case of fading multipath channels, a MIMOconfiguration may increase system capacity by nearly M additionalbits/cycle for each 3-dB increase in SNR.

However, the widespread deployment of multi-antenna systems in wirelesscommunications, particularly in wireless handset devices, has beenlimited by the increased cost that results from increased size,complexity, and power consumption. Providing a separate RF chain foreach transmit and receive antenna is a direct factor that increases thecost of multi-antenna systems. Each RF chain generally comprises a lownoise amplifier (LNA), a filter, a downconverter, and ananalog-to-digital converter (ADC). In certain existing single-antennawireless receivers, the single required RF chain may account for over30% of the receiver's total cost. It is therefore apparent that as thenumber of transmit and receive antennas increases, the systemcomplexity, power consumption, and overall cost may increase. This posesproblems for mobile system designs and applications.

Further limitations and disadvantages of conventional and traditionalapproaches will become apparent to one of skill in the art, throughcomparison of such systems with some aspects of the present invention asset forth in the remainder of the present application with reference tothe drawings.

BRIEF SUMMARY OF THE INVENTION

A system and/or method for single weight antenna system for HSDPA,substantially as shown in and/or described in connection with at leastone of the figures, as set forth more completely in the claims.

Various advantages, aspects and novel features of the present invention,as well as details of an illustrated embodiment thereof, will be morefully understood from the following description and drawings.

BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS

FIG. 1 a is a technology timeline indicating evolution of existing WCDMAspecification to provide increased downlink throughput.

FIG. 1 b illustrates an exemplary High Speed Downlink Packet Access(HSDPA) distributed architecture that achieves low delay linkadaptation, in connection with an embodiment of the invention.

FIG. 1 c illustrates an exemplary Layer 1 HARQ control situated in abase station to remove retransmission-related scheduling and storingfrom the radio network controller, in connection with an embodiment ofthe invention.

FIG. 1 d is a chart illustrating exemplary average carried loads forHSDPA-based macrocell and microcell systems, in connection with anembodiment of the invention.

FIG. 2 is a block diagram of exemplary mobile receiver front-end, inaccordance with an embodiment of the invention.

FIG. 3 is diagram of exemplary HSDPA transmit time intervals, inaccordance with an embodiment of the invention.

FIG. 4 is diagram of exemplary phase control signal, in accordance withan embodiment of the invention.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method and systemfor single weight antenna system for HSDPA. Aspects of method and systemmay comprise receiving HSDPA signals via a plurality of receive antennasand individually adjusting a phase of a portion of the received HSDPAsignals via a single weight. The phase adjusted portion of the receivedHSDPA signals may be combined with at least one of the received HSDPAsignals to generate combined signals. At least one control signal maycontrol the adjusting of the phase of the portion of the received HSDPAsignals.

The single weight that may be used to adjust the phase of a portion ofthe received HSDPA signals may be determined at an identified time. Aplurality of the discrete phases may be communicated to a phase shiftadjuster and utilized to adjust the phase of the portion of the receivedHSDPA signals. The plurality of discrete phases may within a range fromzero radians to substantially 2π radians. Phase shift channel estimatesmay be generated during the identified time to determine the discretephase. A desired phase may be generated from the phase shift channelestimates, and the single weight may be generated from the desiredphase.

FIG. 1 b illustrates an exemplary HSDPA distributed architecture thatachieves low delay link adaptation, in connection with an embodiment ofthe invention. Referring to FIG. 1 b, there is shown terminals 110 and112 and a base station (BS) 114. HSDPA is built on a distributedarchitecture that achieves low delay link adaptation by placing keyprocessing at the BS 114 and thus closer to the air interface asillustrated. HSDPA leverages methods that are well established withinexisting GSM/EDGE standards, including fast physical layer (L1)retransmission combining and link adaptation techniques, to deliversignificantly improved packet data throughput performance between themobile terminals 110 and 112 and the BS 114.

The HSDPA technology employs several important new technologicaladvances. Some of these may comprise scheduling for the downlink packetdata operation at the BS 114, higher order modulation, adaptivemodulation and coding, hybrid automatic repeat request (HARQ), physicallayer feedback of the instantaneous channel condition, and a newtransport channel type known as high-speed downlink shared channel(HS-DSCH) that allows several users to share the air interface channel.When deployed, HSDPA may co-exist on the same carrier as the currentWCDMA and UMTS services, allowing operators to introduce greatercapacity and higher data speeds into existing WCDMA networks. HSDPAreplaces the basic features of WCDMA, such as variable spreading factorand fast power control, with adaptive modulation and coding, extensivemulticode operation, and fast and spectrally efficient retransmissionstrategies.

In current-generation WCDMA networks, power control dynamics are on theorder of 20 dB in the downlink and 70 dB in the uplink. WCDMA downlinkpower control dynamics are limited by potential interference betweenusers on parallel code channels and by the nature of WCDMA base stationimplementations. For WCDMA users close to the base station, powercontrol cannot reduce power optimally, and reducing power beyond the 20dB may therefore have only a marginal impact on capacity. HSDPA, forexample, utilizes advanced link adaptation and adaptive modulation andcoding (AMC) to ensure all users enjoy the highest possible data rate.AMC therefore adapts the modulation scheme and coding to the quality ofthe appropriate radio link.

FIG. 1 c illustrates an exemplary Layer 1 HARQ control situated in abase station to remove retransmission-related scheduling and storingfrom the radio network controller, in connection with an embodiment ofthe invention. Referring to FIG. 1 c, there is shown a hybrid automaticrepeat request (HARQ) operation, which is an operation designed toreduce the delay and increase the efficiency of retransmissions. Layer 1HARQ control is situated in the Node B, or base station (BS), 122 thusremoving retransmission-related scheduling and storing from the radionetwork controller (RNC) 120. This HARQ approach avoids hub delay andmeasurably reduces the resulting retransmission delay.

For example, when a link error occurs, due to signal interference orother causes, a mobile terminal 124 may request the retransmission ofthe data packets. While current-generation WCDMA networks handle thoseretransmission requests through the radio network controller 120, HSDPAretransmission requests are managed at the base station 122.Furthermore, received packets are combined at the physical (PHY) layerand retrieved only if successfully decoded. If decoding has failed, thenew transmission is combined with the old transmission before channeldecoding. The HSDPA approach allows previously transmitted frames (thatfailed to be decoded) to be combined with the retransmission. Thiscombining strategy provides improved decoding efficiencies and diversitygains while minimizing the need for additional repeat requests.

While the spreading factor may be fixed, the coding rate may varybetween ¼ and ¾, and the HSDPA specification supports the use of up to10 multicodes. More robust coding, fast HARQ, and multi-code operationeliminates the need for variable spreading factor and also allows formore advanced receiver structures in the mobile such as equalizers asopposed to the traditional RAKE receiver used in most CDMA systems. Thisapproach may also allow users having good signal quality or highercoding rates and those at the more distant edge of the cell having lowercoding rates to each receive an optimum available data rate.

By moving data traffic scheduling to the base station 122, and thuscloser to the air interface, and by using information about channelquality, terminal capabilities, QoS, and power/code availability, HSDPAmay achieve more efficient scheduling of data packet transmissions.Moving these intelligent network operations to the base station 122allows the system to take full advantage of short-term variations, andthus to speed and simplify the critical transmission scheduling process.The HSDPA approach may, for example, manage scheduling to track the fastfading of the users and when conditions are favorable to allocate mostof the cell capacity to a single user for a very short period of time.At the base station 122, HSDPA gathers and utilizes estimates of thechannel quality of each active user. This feedback provides currentinformation on a wide range of channel physical layer conditions,including power control, ACK/NACK ratio, QoS, and HSDPA-specific userfeedback.

While WCDMA Release 99 or WCDMA Release 4 may support a downlink channel(DCH) or a downlink shared channel (DSCH), the HSDPA operation providedby WCDMA Release 5 may be carried on a high-speed downlink sharedchannel (HS-DSCH). This higher-speed approach uses a 2-ms interval framelength (also known as time transmit interval), compared to DSCH framelengths of 10, 20, 40 or 80 ms. DSCH utilizes a variable spreadingfactor of 4 to 256 chips while HS-DSCH may utilize a fixed spreadingfactor of 16 with a maximum of 15 codes. HS-DSCH may support 16-levelquadrature amplitude modulation (16-QAM), link adaptation, and thecombining of retransmissions at the physical layer with HARQ. HSDPA alsoleverages a high-speed shared control channel (HS-SCCH) to carry therequired modulation and retransmission information. An uplink high-speeddedicated physical control channel (HS-DPCCH) carries ARQacknowledgements, downlink quality feedback and other necessary controlinformation on the uplink.

FIG. 1 d is a chart illustrating exemplary average carried loads forHSDPA-based macrocell and microcell systems, in connection with anembodiment of the invention. Referring to chart 130 in FIG. 1 d, inpractical deployments, HSDPA more than doubles the achievable peak userbit rates compared to WCDMA Release 99. With bit rates that arecomparable to DSL modem rates, HS-DSCH may deliver user bit rates inlarge macrocell environments exceeding 1 Mbit/s, and rates in smallmicrocells up to 5 Mbit/s. The HSDPA approach supports bothnon-real-time UMTS QoS classes and real-time UMTS QoS classes withguaranteed bit rates.

Cell throughput, defined as the total number of bits per secondtransmitted to users through a single cell, increases 100% with HSDPAwhen compared to the WCDMA Release 99. This is because HSDPA's use ofHARQ combines packet retransmission with the earlier transmission, andthus no transmissions are wasted. Higher order modulation schemes, suchas 16-QAM, enable higher bit rates than QPSK-only modulation in WCDMARelease 99, even when the same orthogonal codes are used in bothsystems. The highest throughput may be obtained with low inter-pathinterference and low inter-cell interference conditions. In microcelldesigns, for example, the HS-DSCH may support up to 5 Mbit/s per sectorper carrier, or 1 bit/s/Hz/cell.

FIG. 2 is a block diagram of exemplary mobile receiver front-end, inaccordance with an embodiment of the invention. Referring to FIG. 2,there is shown a transmitter section 200 a, a receiver section 200 b, aprocessor 200 c, and a memory block 200 d. The transmitter section 200 amay comprise a base station 202 and transmit antennas 204 a and 204 b.The receiver section 200 b may comprise receiver antennas 205 a and 205b, RF bandpass filters (BPF) 206 and 212, low-noise amplifiers (LNA) 208and 214, a phase shift adjuster (PSA) 216, a RF combiner 218, and a RFblock 220. The receiver section 200 b further comprises a chip matchedfilter (CMF) block 230, a cluster path processor (CPP) block 240, and asingle weight baseband generator (SWBBG) block 250. The SWBBG block 250may comprise a single weight channel estimator block 254, a singleweight algorithm block 252, and a phase rotation block 256.

The base station 202 in the transmitter section 200 a may transmit datat_(x1) and t_(x2) via the transmit antennas 204 a and 204 b,respectively. The transmitted data from the transmit antenna 204 a mayhave propagation paths to the receiver antennas 205 a and 205 b that mayhave an aggregate time varying impulse response of h ₁₁ and h ₁₂,respectively. Similarly, the transmitted data from the transmit antenna204 b may have propagation paths to the receive antennas 205 a and 205 bthat may have an aggregate time varying impulse response of h ₂₁ and h₂₂, respectively. The time varying impulse responses that correspond tothe propagation paths of the signals received by each of the receiveantennas 205 a and 205 b may be represented by the channel responses h ₁and h ₂, respectively. These channel responses may be modeled as thealgebraic sum of the time varying impulse response h ₁₁ and h ₂₁ for thereceive antenna 205 a and h ₁₂ and h ₂₂ for the receive antenna 205 b.

The receiver section 200 b may comprise suitable logic, circuitry,and/or code that may be adapted to receive RF signals, process the RFsignals by filtering, amplifying and/or adjusting phase and/or amplitudeof the RF signals, and converting the processed RF signals to digitalbaseband signals. Furthermore, the receiver section 200 b may be adaptedto generate a single weight control signal based on processed digitalbaseband signals. Specifically, the BPFs 206 and 212 may comprisesuitable logic and/or circuitry that may be adapted to receive a RFinput, limit the frequencies of the signal to a determined band offrequencies, and output that band of frequencies. The LNAs 208 and 214may comprise suitable logic and/or circuitry that may be adapted toreceive an input signal, and amplify the input signal while introducingvery little additional noise.

The PSA 216 may comprise suitable logic and/or circuitry that may beadapted to receive a control signal that may stimulate a change in thephase of an RF input signal. The phase change may be in exemplaryincrements of kΔφ, where k may be an integer variable and Δφ may be aminimal phase change allowed. The minimal phase change Δφ may be designand/or implementation dependent. An embodiment of the invention mayallow the variable k to range from 0 to N−1, where N*(Δφ) may be 2π.

The RF combiner 218 may comprise suitable logic and/or circuitry thatmay be adapted to have as inputs a plurality of analog RF signals andoutput a combined analog RF signal that may be a sum of the plurality ofanalog RF signals. The RF block 220 may comprise suitable logic,circuitry, and/or code that may be adapted to receive an analog RFsignal, and amplify, filter and/or otherwise convert the analog RFsignal to a digital baseband signal for further processing. The CMFblock 230 may comprise suitable logic, circuitry and/or code that may beadapted to digitally filter the digital baseband signal for a WCDMAbaseband bandwidth. The CMF block 230 may comprise a plurality ofdigital filters for an in-phase (I) component and a quadrature (Q)component of the digital baseband signal. The digital filters may have acombined impulse response that may be square root raised cosine (SRRC),which may be required by the WCDMA specifications.

The CPP block 240 may comprise suitable logic, circuitry, and/or codethat may be adapted to track time-wise clusters of multipath signals andto estimate the complex phase and/or amplitude of the multipath signalsin the signal clusters. The signal cluster may comprise an aggregate ofreceived signals with maximum time difference that may be no more than16/(3.84×10⁶) seconds. U.S. application Ser. No. ______ (Attorney DocketNo. 16218US02) provides a detailed description of signal clusters and ishereby incorporated herein by reference in its entirety. The CPP block240 may be adapted to determine channel estimates ĥ ₁ and ĥ ₂ of thetime varying impulse responses of the channels, for example, themultipath vectors h ₁ and h ₂. The CPP 240 may output the estimates as:

{circumflex over (h)}₁+({circumflex over (h)}₂*e^(jφ)).

The complex notation e^(jφ) may describe a pair of functions cos(φ) andsin(φ), where j may be the square root of −1. Therefore, e^(jφ) maydescribe a change of RF phase of φ that may be associated with thechannel response {circumflex over (h)}₂. The single weight channelestimator block 254 may comprise suitable logic, circuitry, and/or codethat may be adapted to receive the channel estimates from the CPP 240during a RF phase learning stage. The RF phase learning stage may takeplace during a defined period for collecting signals that is used togenerate a new phase φ₀ for the PSA 216. The RF phase learning stage maybe indicated by a control signal from the processor 200 c, for example.The channel estimates may be processed, and at the end of the RF phaselearning stage, the single weight channel estimator block 254 may outputthe channel estimates {circumflex over (h)}₁ and {circumflex over (h)}₂.The channel estimates ĥ₁ and {circumflex over (h)}₂ may be calculatedusing the following equations:

ĥ _(1,i)=(N)^(−1/2)Σ_(k=0, N−1) CCPOutput(k)_(i)

ĥ _(1,i)=(N)^(−1/2)Σ_(k=0, N−1) CCPOutput(k)_(i)exp(jkΔφ)_(i)

The CPPOutput(k) may be an output from the CPP 240 during the RF phaselearning stage, and may be expressed as:

CCPOutput(k)_(i) =ĥ _(1,i)+(ĥ ₂*exp(jkΔφ)_(i))

The variable k may be similar to the variable k used with respect to thedescription of the PSA 216. The variable i may indicate a multipathreceived by a receive antenna.

The single weight algorithm block 252 may comprise suitable logic,circuitry, and/or code that may be adapted to receive the phase shiftchannel estimates {circumflex over (h)}₁ and {circumflex over (h)}₂ togenerate a phase φ_(r). The phase φ_(r) may be the phase of z, where zmay be defined as:

z=Σ _(i=0, L−1)(ĥ* _(2,i))(ĥ _(1,i)),

φ_(r)=Phase{z}

where {circumflex over (h)}*_(2,i) may be a complex conjugate of{circumflex over (h)}_(2,i). The phase φ_(r) may represent a phasecorrection and may be communicated to the phase rotation block 256.

The phase rotation block 256 may comprise suitable logic, circuitry,and/or code that may be adapted to receive an asserted rotation commandfrom a processor, for example, the processor 200 c, during the RF phaselearning stage, and other stages as may be necessary. The phase rotationblock 256 may communicate via a control signal to the PSA 216 during theRF phase learning stage a plurality of phases from 0 to 2π radians.Therefore, the PSA 216 may adjust the phase of the received signal fromthe receive antenna 205 b (FIG. 2) by the plurality of phases from 0 to2π. The plurality of phase changes may be indicated by exp(jkΔφ) where kmay range from 0 to N−1, such that NΔφ is equal to 2π. The duration ofthe RF phase learning stage may be determined by a period T, where T maybe the duration during which each value k may be asserted. This may befurther illustrated in FIG. 4.

Additionally, the single weight channel estimator block 254 may use theoutput φ_(r) of the single weight algorithm block 252 to calculate thenew phase φ₀. In an embodiment of the invention, the phase rotationblock 256 may determine k₀, where k₀ may be the value of k that mayproduce the desired phase φ₀ that may be closest to the phase φ_(r)communicated by the single weight algorithm block 252. The variable k₀may be determined by using the following equation:

k ₀=Minimum(|φ_(r)−φ(k)|)_(k=0, N−1)

After determining the variable k₀, the phase φ₀ may be determined as:

φ₀=k₀Δφ.

The result of this process may be that the phase φ₀ may be generatedusing the control signal k₀, where k₀ may be a value from the set {k:k=0, . . . , N−1} The SWBBG 250 may, therefore, be commanded either togenerate the plurality of phases from 0 to 2π using the set of functionsignal exp(jkΔφ), or the desired phase through the pair of values:exp(jk0Δφ). For example, the values {exp(jkΔφ), k=0, . . . , N−1} may bestored in a lookup-table such that the value of k may be the addressassociated with the function pair {Cos(kΔφ), Sin(kΔφ)} that are stored.The PSA 216 may, for example, use the conversion table 217 a to convertthe pair of numeric values that may be received into a numeric phase.The PSA 216 may also use the D/A converter 217 b to convert the numericphase into an analog value.

Although the control signal from the SWBBG 250 to the PSA 216 may havebeen described as being looked up in a look-up table 257, the inventionneed not be so limited. For example, the lookup table 257 and aconversion table 217 a may be in different physical locations, or theymay be part of the same memory block, for example, the memory block 200d.

The processor 200 c may comprise suitable logic, circuitry, and/or codethat may be adapted to monitor and/or control various functionalities ofa mobile terminal. For example, the processor 200 c may be adapted tomonitor the rate of change in the measured channel response {circumflexover (h)}₁+({circumflex over (h)}₂*e^(jφ)) generated by the CPP 240 andgenerate an estimate of the moving speed of the mobile terminal. Basedon the estimate of the mobile terminal moving speed, the processor 200 cmay determine how often the SWBBG 250 may perform a new phase φ₀ for thePSA 216. The processor 200 c may communicate a control command to theSWBBG 250. The control command may indicate to the SWBBG 250 whether toenter the RF phase learning stage to determine a new phase φ0 that maybe communicated to the PSA 216.

Although the processor 200 c may have been described as communicating acommand to the SWBBG 250 to enter the RF phase learning stage, theinvention need not be so limited. For example, a hardware circuit may beused to monitor, data reception in order to determine when and/or howoften the RF phase learning stage may take place.

The memory block 200 d may be used to store code and/or data, and may bea writeable medium, for example, RAM. Portions of the memory block 200 dmay comprise look-up tables and/or conversion tables.

FIG. 3 is a diagram of exemplary HSDPA transmit time intervals, inaccordance with an embodiment of the invention. Referring to FIG. 3,there is shown a signal receive diagram 310, a first exemplary RF phaselearning stage diagram 320, and a second exemplary RF phase learningstage diagram 330. The signal receive diagram 310 may show threetransmit time intervals (TTIs) 312, 314 and 316 that may be utilized byHSDPA transmitters, for example, the base station 202 (FIG. 2), totransmit data packets. The TTIs 312, 314, and 316 may start at timeinstances t₀, t₂, and t₄, respectively, and may end at times t₂, t₄, andt₆, respectively. Each TTI may be 2 milliseconds (mS) in duration. Amobile terminal may be shown as receiving packet data during the TTIs312 and 316, while not receiving any packet data during the TTI 314.However, common channels, in particular, a phase reference channel knownas CPICH is always transmitted if the mobile terminal is in a HSDPAnetwork.

The mobile terminal may determine a phase φ₀ via the RF phase learningstage at the beginning of any TTI. However, if the RF phase learningstage occurs in a TTI when packet data is being received, a portion ofthe packet data may be in unfavorable receiving condition. Therefore, ifpossible, it may be useful to generate the phase φ₀ during the TTIs whenno packet data is being received. The first exemplary RF phase learningstage diagram 320 may illustrate a case where the RF phase learningstage occurs during a TTI, for example, the TTI 314, when no packet databeing received. The RF phase learning stage may start at the beginningof the TTI 314 at time instant t₂, and end at time instant t₃. The phaseφ₀ generated during the TTI 314 may be used for future TTIs, forexample, the TTI 316. How often a new phase φ₀ is generated may bedesign and/or implementation dependent.

Although generating the phase φ0 during TTIs when no packet data isbeing received may be useful when the mobile terminal is moving slowlyor not at all, the phase φ0 may have to be generated more frequentlywhen the mobile terminal is moving faster. The determination of howoften and when to generate the phase φ0 may be design and/orimplementation dependent. An embodiment of the invention may generatethe phase φ0 so that it may be used during the TTI when packet data arebeing received, for example, the TTIs 312 and 316. The second exemplaryRF phase learning stage diagram 320 may illustrate this. The RF phaselearning stages may start at time instances t0 and t4, and end at timeinstances t1 and t5. The phase φ0 may be used during the TTI when it wasgenerated. For example, the phase φ0 generated in TTI 312 may be usedduring the TTI 312 to receive packet data, and the phase 40 generated inTTI 316 may be used during the TTI 316. Although the RF phase learningstage may induce an unfavorable condition for a short period for signalreception, the benefit from using the RF phase learning stage mayoutweigh the non-optimal two antenna signal-combining that may occurotherwise.

FIG. 4 is a diagram of exemplary phase control signal, in accordancewith an embodiment of the invention. Referring to FIG. 4, there is showna diagram that illustrates the control signal that may be generated bythe phase rotation block 256 during the RF phase learning stage. Thecontrol signal may comprise phase by which the PSA 216 (FIG. 2) maychange the phase of the input signal received by the receive antenna 205b. There may be N distinct discrete phases 402, 404, 406, . . . , 408,communicated by the control signal, and each phase may last for a periodT. Each discrete phase may be communicated by the signal exp(jkΔφ),where k may range from 0 to N−1. Therefore, a plurality of phases 402,404, 406, . . . , 408, ranging from 0 to exp(j((N−1)/N)Δφ) may becommunicated to the PSA 216. The time N*T may be determined to be theperiod of the RF phase learning stage. N and/or T may be design and/orimplementation dependent.

Although the sequence described here is based on HSDPA communicationnetwork, it will be obvious that the invention can be utilized in manyother type of communication networks

Aspects of the system may comprise a plurality of receive antennas 205 aand 205 b (FIG. 2) that receives HSDPA signals. A phase shift adjuster216 (FIG. 2) may individually adjust a phase of a portion of thereceived HSDPA signals via a single weight. A RF combiner 218 (FIG. 2)may combine the phase adjusted portion of the received HSDPA signalswith at least one of the received signals to generate combined HSDPAsignals. A single weight baseband generator 250 (FIG. 2) may generate atleast one control signal that comprises the single weight that controlsthe adjusting of the phase of the portion of the received HSDPA signals.The single weight baseband generator 250 may determine a discrete phaseto phase adjust a portion of the received HSDPA signals. A processor 200c (FIG. 2) may identify a time to determine a discrete phase to adjust aportion of the received HSDPA signals and generate appropriate controlssignals.

The single weight baseband generator 250 may communicate a plurality ofthe discrete phases 402, 404, 406, . . . , 408 (FIG. 4) to adjust thephase of the portion of the received HSDPA signals, where the pluralityof the discrete phases 402, 404, 406, . . . , 408 may range from zeroradians to substantially 2π radians. A single weight channel estimator254 (FIG. 2) may receive a channel estimate from the CPP block 240 andgenerate channel estimates for the receiver antennas 205 a and 205 b. Asingle weight algorithm block 252 (FIG. 2) may generate a desired phasebased on the channel estimate for antennas 205 a and 205 b. A phaserotation block 256 (FIG. 2) may generate the single weight thatconstitutes the desired RF phase.

Accordingly, the present invention may be realized in hardware,software, or a combination of hardware and software. The presentinvention may be realized in a centralized fashion in at least onecomputer system, or in a distributed fashion where different elementsare spread across several interconnected computer systems. Any kind ofcomputer system or other apparatus adapted for carrying out the methodsdescribed herein is suited. A typical combination of hardware andsoftware may be a general-purpose computer system with a computerprogram that, when being loaded and executed, controls the computersystem such that it carries out the methods described herein.

The present invention may also be embedded in a computer programproduct, which comprises all the features enabling the implementation ofthe methods described herein, and which when loaded in a computer systemis able to carry out these methods. Computer program in the presentcontext means any expression, in any language, code or notation, of aset of instructions intended to cause a system having an informationprocessing capability to perform a particular function either directlyor after either or both of the following: a) conversion to anotherlanguage, code or notation; b) reproduction in a different materialform.

While the present invention has been described with reference to certainembodiments, it will be understood by those skilled in the art thatvarious changes may be made and equivalents may be substituted withoutdeparting from the scope of the present invention. In addition, manymodifications may be made to adapt a particular situation or material tothe teachings of the present invention without departing from its scope.Therefore, it is intended that the present invention not be limited tothe particular embodiment disclosed, but that the present invention willinclude all embodiments falling within the scope of the appended claims.

1-24. (canceled)
 25. A MIMO phone having at least a first antenna and asecond antenna, comprising: a first bandpass filter operatively coupledto a first low noise amplifier, wherein the first bandpass filter isoperatively disposed between the first low noise amplifier and the firstantenna; a second bandpass filter operatively coupled to a second lownoise amplifier, wherein the second bandpass filter is operativelydisposed between the second low noise amplifier and the second antenna;a phase shift adjuster operatively coupled to an RF combiner, whereinthe phase shift adjuster is operatively disposed between the RF combinerand the second low noise amplifier, wherein the RF combiner combines afirst RF analog signal output from the first low noise amplifier and asecond RF analog signal output from the phase shift adjuster; a firstprocessor operatively coupled to the RF combiner, wherein the firstprocessor is configured to track time-wise clusters of multipath signalsand to estimate a complex phase or an amplitude of the multipath signalsin the signal clusters; and a baseband weight generator that receivesinformation from the first processor and generates a phase controlsignal to the phase shift adjuster.
 26. The MIMO phone according toclaim 25, wherein the MIMO phone supports HSDPA.
 27. The MIMO phoneaccording to claim 26, wherein the HSDPA co-exists on a same carrier asWCDMA and UMTS services.
 28. The MIMO phone according to claim 25,wherein the MIMO phone supports one or more of the following: adaptivemodulation, adaptive coding, multicode operation and retransmissionstrategies.
 29. The MIMO phone according to claim 25, comprising: a chipmatched filter operatively coupled to the first processor, wherein thechip matched filter comprises a plurality of digital filters for anin-phase component and a quadrature component of a baseband signal. 30.The MIMO phone according to claim 29, wherein the chip matched filter isconfigured to digitally filter the baseband filter for a WCDMA basebandbandwidth.
 31. The MIMO phone according to claim 25, wherein the phaseshift adjuster is configured to adjust a phase of a signal received fromthe second low noise amplifier in discrete increments.
 32. The MIMOphone according to claim 31, wherein the discrete increments are integermultiples of a minimum phase change.
 33. The MIMO phone according toclaim 25, wherein the phase shift adjuster comprises one or more of thefollowing: a digital-to-analog converter and a conversion table.
 34. TheMIMO phone according to claim 33, wherein the phase shift adjuster usesthe conversion table to convert a pair of numeric values into a numericphase.
 35. The MIMO phone according to claim 25, wherein the basebandweight generator comprises a phase rotator and a lookup table, whereinthe lookup table provides the phase control signal to the phase shiftadjuster.
 36. The MIMO phone according to claim 25, comprising a secondprocessor operatively coupled to the baseband weight generator, whereinthe second processor monitors or controls particular functionalities ofthe MIMO phone.
 37. The MIMO phone according to claim 36, wherein thefirst processor generates a measured channel response, and wherein thesecond processor monitors a rate of change in the measured channelresponse to estimate a moving speed of the MIMO phone.
 38. The MIMOphone according to claim 37, wherein the second processor determines howoften the baseband weight generator provides a new phase control signalto the phase shift adjuster based on the estimated moving speed of theMIMO phone.
 39. The MIMO phone according to claim 25, comprising RFcircuitry operatively coupled to the RF combiner, wherein the RFcircuitry is operatively disposed between the RF combiner and theprocessor.
 40. The MIMO phone according to claim 39, wherein the RFcircuitry is configured to amplify, to filter and to convert a combinedRF signal received from the RF combiner into a combined baseband signal.41. The MIMO phone according to claim 25, wherein the first bandpassfilter, the second bandpass filter, the first low noise amplifier, thesecond bandpass filter, the phase shift adjuster and the RF combiner arepart of a MIMO receiver.
 42. The MIMO phone according to claim 25,wherein the MIMO phone supports one or more of the following: WCDMA,GPRS, EDGE and UMTS.
 43. The MIMO phone according to claim 25, whereinthe MIMO phone supports Layer 1 HARQ control.
 44. The MIMO phoneaccording to claim 25, wherein the MIMO phone supports using up to tenmulticodes.